An output signal from a power amplifier for wireless transmission generally has a limitation on linearity, and in particular, a gain is decreased when a level of an input signal is large (linearity distortion). As a circuit for compensating such a linearity distortion, a Cartesian feedback type distortion correction apparatus is known. If the Cartesian feedback type distortion correction apparatus functions ideally, a high linearity may be obtained in the output signal from the power amplifier.
In the Cartesian feedback type distortion correction apparatus, the output signal from the power amplifier is taken out and fed back to an input side. At this time, for example, a phase change in a feed back system is generated along with an influence of an antenna load, a propagation delay of a directional coupler or a demodulator, or the like. Therefore, in order to effectively operate the Cartesian feedback type distortion correction apparatus, it is necessary to correct this phase change in the feed back system.
From the above-mentioned viewpoint, a phase correction apparatus applied to the Cartesian feedback type distortion correction apparatus is known. FIG. 1 illustrates a main part of this phase correction apparatus.
In FIG. 1, an in-phase component I and a quadrature-phase component Q of a baseband signal for transmission are combined after being modulated by a quadrature-phase modulator 40. This combined signal is amplified by a power amplifier 90 (PA) to a desired level and transmitted as an RF signal (RF_OUT). Also, a part of this RF signal (RF_OUT) is taken out (feed back) by a directional coupler, and a baseband signal (an in-phase component I*, a quadrature-phase component Q*) is generated by a quadrature-phase demodulator 30 from a feedback signal y(t). Herein, based on the above-mentioned phase change in the feed back system (in FIG. 1, a delay equivalent to the phase change is denoted as delay period τ), I≠I* and Q≠Q* are established, and in order to correct this phase change, a phase correction apparatus is provided.
In FIG. 1, the phase correction apparatus is provided with a phase detector including a sine detection unit 101 and a cosine detection unit 102, and a phase shifter 104. Herein, when a target correction amount of the phase (that is, a phase error) is set as Δφ, the fed-back baseband signal (the in-phase component I*, the quadrature-phase component Q*) is represented as in the following Expression (1) and Expression (2). In Expression (1) and Expression (2), only when Δφ=0, I=I* and Q=Q* are established. Also, sin(Δφ) is calculated by the sine detection unit 101 based on the following Expression (3). In Expression (3), k denotes a normalized constant, and k=1/(I·I+Q·Q).I*=I·cos(Δφ)+Q·sin(−Δφ)  (1)Q*=I·sin(Δφ)+Q·cos(Δφ)  (2)sin(Δφ)=k·(I·Q*−Q·I*)  (3)
Also, the cosine detection unit 102 calculates cos(Δφ) while following a relation of cos(Δφ)=(1−sin2(Δφ))1/2. In the cosine detection unit 102, a configuration is adopted in which a square sum of the input to the phase shifter 104 becomes a given constant Mag, and with this configuration, the compensation is made so that the amplitude of the output signal from the phase shifter 104 is constant.
As illustrated in the following Expression (4), the phase shifter 104 respectively multiplies a carrier signal sin(ωt) from a local oscillator and a signal obtained by shifting the carrier signal by π/2 by cos(Δφ) and sin(Δφ) to be combined. As a result, a signal sin(ωt+Δφ) in which the phase is advanced by Δφ with respect to the carrier signal sin(ωt) from the local oscillator is supplied to the quadrature-phase modulator 40. Therefore, the phase error between the RF signal y(t) fed back from the output of the wireless transmitter and the carrier signal provided to the quadrature-phase modulator 40 becomes 0 (Δφ=0).cos(Δφ)·sin(ωt)+sin(Δφ)·cos(ωt)=sin(ω·t+Δφ)  (4)
Herein, the phase correction apparatus described with reference to FIG. 1 may basically perform the correction when the phase error Δφ is in a range of −π/2<Δφ<+π/2. In view of the above, a phase correction apparatus is desired which may perform the correction in a range of the entire phase range, that is, −π<Δφ<+π.
As the phase correction apparatus which may perform the correction in the range of the entire phase range, an apparatus which is provided with a phase shifter for shifting the phase of the fed-back baseband signal in π/2 unit is known. Hereinafter, a description will be given of this phase correction apparatus with reference to FIG. 2.
In FIG. 2, the phase shifter 104 may perform the correction in a range of −π/2<Δφ<+π/2. Also, the phase shifter 104 is provided with a π/2 phase shifter 108 capable of shifting the fed-back baseband signal (the in-phase component I*, the quadrature-phase component Q*) from the quadrature-phase demodulator 30 in the π/2 unit. The phase detector 103 detects the phase error (the target correction amount) from the baseband signal for transmission (the in-phase component I, the quadrature-phase component Q) and the fed-back baseband signal (the in-phase component I*, the quadrature-phase component Q*). Then, in a case where the target correction amount exceeds the range of −π/2<Δφ<+π/2, the phase detector 103 sends out a signal for shifting the phase of the fed-back baseband signal (the in-phase component I*, the quadrature-phase component Q*) in the π/2 unit to the π/2 phase shifter 108.
Incidentally, in the phase correction apparatus illustrated in FIG. 2, a transfer function H(s) of the control system (s: Laplace operator) is as illustrated in Expression (5). In Expression (5), the setting is as follows.
K(s): A transfer function on a direct line between the input and output of the system
α(s): A transfer function of the π/2 phase shifter 108
β(s): A transfer function on a feed back line except for the π/2 phase shifter 108
                              H          ⁡                      (            s            )                          =                                            K              ⁡                              (                s                )                                                    1              +                                                β                  ⁡                                      (                    s                    )                                                  ·                                  α                  ⁡                                      (                    s                    )                                                  ·                                  K                  ⁡                                      (                    s                    )                                                                                ≈                      1                                          β                ⁡                                  (                  s                  )                                            ·                              α                ⁡                                  (                  s                  )                                                                                        (        5        )            
Expression (5) means that the transfer function H(s) of the control system depends on the transfer functions α(s) and β(s) on the feed back line. Therefore, in the phase correction apparatus illustrated in FIG. 2, a characteristic of the output of the control system (that is, the output of the power amplifier) is easily affected by a non-linearity of the π/2 phase shifter 108 and a distortion characteristic. This situation is not preferable in terms of a stable function of the phase correction operation.